Polyphase motor system with dc motor characteristics

ABSTRACT

An electric induction motor/control system that provides the desirable high starting torque and wide speed characteristics of direct current motors. The stator of the induction motor has two specially interrelated winding sets per phase. The motor windings are supplied with unidirectional current pulses from silicon controlled rectifiers that are programmed to provide advantageous modes of operation for the motor. The unique relation and interaction between the motor windings hereof and the associated electronic circuitry results in high power factor, good efficiency, and self-clearing SCR commutation action. The preferred motors use a squirrel cage rotor with no commutator, brushes, or slip rings. The complementary stator winding arrangements are not voltage limited, and thereby permit the construction of large power motor systems. Practical motor ratings are from fractional to over 500 horsepower, at speeds ranging from 600 to 40,000 rpm, and higher. The electronic control sections for even the high horsepower motors are operated at low voltage and with few watts. Control circuitry is provided that can operate the motor systems: (a) at constant output torque over a wide speed range; (b) at constant horsepower over a selected speed range; (c) with traction output characteristics similar to that of series motors; (d) at constant speed; (e) and with direct speed reversal in any of these modes. These motor systems require little maintenance, and may be hermetically sealed.

United States Patent [191 Greenwell POLYPHASE MOTOR SYSTEM WITH DC MOTORCHARACTERISTICS .IHWPKQEZJQQE: Gre nwq iR n [73] Assignee: Lear MotorsCorporatiom'Reno,

Filed: May 19, 1971 Appl. No.: 144,897

11/1971 Graham 318/225 R X 6/1965 Sisk et al 318/225 R PrimaryExaminer-Gene Z Rubinson Attorney-Richard A. Marsen ABSTRACT An electricinduction motor/control system that provides the desirable high startingtorque and wide speed characteristics of direct current motors. Thestator of Aug. 14, 1973 the induction motor has two speciallyinterrelated winding sets per phase. The motor windings are suppliedwith unidirectional current pulses from silicon controlled rectifiersthat are programmed to provide advantageous modes of operation for themotor. The unique relation and interaction between the motor windingshereof and the associated electronic circuitry results in high powerfactor, good efficiency, and selfclearing SCR commutation action. Thepreferred motors use a squirrel cage rotor with no commutator, brushes,or slip rings. The complementary stator winding arrangements arenotvoltage limited, and thereby permit the construction of large powermotor systems. Practical motor ratings are from fractional to over 500horsepower, at speeds ranging from 600 to 40,000 rpm, and higher. Theelectronic control'sections for even the j high horsepower motors areoperated at low voltage and with few watts. Control circuitry isprovided that can operate the motor systems: (a) at constant outputtorque over a wide speedrange; (b) at constant horsepower over aselected speed range; (c) with traction output characteristics similarto that of series motors; I

(d) at constant speed; (e) and with direct speed reversal in any ofthese modes. These motor systems require little maintenance, and may behermetically sealed.

13 Claims, 21 Drawing Figures ta l 4i 32-I 2-4 W 2 SCR GATE DRIVER ANDSEQUENTIAL TRIGGERING CIRCUITS (LOGIC) 3 (CONTROL) PMENTEBIIIB 14 msSHEET U 0F 6 73 4 3-PHASE N 4 ""L RECTIFIER 74 /75 7? 79F FIG- SCR:MOTOR 1 CONTROL 1 82 L 83 I2OVDC TORQUE OOVDc (LBS- FT) RATED HP zovocPW O v I I I I I 02 4 e 8|OI2|4|6|82O F o- SHAFT-RPM x I00) TORQUE 0 PMRATED SPEED NOMINAL PEAK TORQUE NORMAL MOTOR REVERSE ROTATION TORQUEPOSITIVE MOTOR ROTATION REGENERATIVE DYNAMIC BRAKING PATEN IEO MIR 14 msSHEET 5 0F 6 FIG. l2

cousmuf HP OUTPUT K C 0 L C IOO SPEED CONTROL TO LOGIC FIG M3010 .ZEFDOSHAFT SPEED \CONSYTANT TORQUE OUTPUT Mm p s R am no A 2 M n m M 8 oUDGmOk FDQFDO n 5 I n 4 n 3 n m w W mm I1 OU fl 0 C I 3 'n G F OUDOKO... .hJnEbO SHAFT SPEED-v FIXED TO LOGIC H 5 K D. m C O m VSA O L:6 UR C l a M F S 0. 5 l J Li G l F BACKGROUND OF THE INVENTION Polyphasealternating current induction motors with squirrel cage rotors are thesimplest of motor drives. However, in conventional use on commercialpower lines their starting torque and speed range are rather limited ascompared to direct current motors. On the other hand, direct currentmotors require commutators and brushes; a source of wear and ofsparking. Electronic solid state circuitry has heretofore been combinedwith induction motors to substantially widen their operatingcharacteristics. Such systems however have been rather unreliable in thefield, and involved considerable circuit complexity, reduced efficiency,and lower power factor. Two general types of such motor systems havebeen evolved. One uses solid state inverters that connect directly tostandard AC motors. Another system utilizes complementary winding setsthat are closely coupled in each phase, such as bifilar windings, thewinding sets being supplied by successive current pulses to operate themotor.

The polyphase motor/inverter system applies square waveform power pulsesto conventional polyphase motors. There are several reasons for itscomparatively poor performance. These polyphase motors, say threephase,are wound with less than full coil pitch to control harmonic content.The short pitch results in some stator slots carrying the applied pulsecurrents in both directions at the same time. This produces less windingeffectiveness when operated on the square-wave power. Further, thepolyphase stators operate at a higher saturation level on such invertedpower, because greater volt-seconds are presented by the squarewaveform. The RMS value of the current supplied from the inverters ishigher than that of fundamental frequency current from a power line.Higher losses thus occurs in the motor windings than when operateddirectly on sine-wave power. Also, the starting currents are higher. Ifthe inverter were thoroughly filtered to supply sinusoidal power at aparticular frequen y, it would cost more and be limited as to its motoroperating polyphase characteristics.

In the bifilar motor winding system the entire phase to phase voltagemust be supported in the stator by the film insulation on the wire.There is no opportunity therein to utilize phase insulation in thestator slots. This severely limits the voltage rating of such motors.Limiting motor voltage rating limits its power output. Further, sincethere is good magnetic linkage between the complementary windings ofeach bifilar set, they effectively cancel each other magnetically andpush the magnetic flux to near zero. As this happens, the phase currentduring commutation is limited mainly by the stator winding resistance,which is low. There is thus a sudden current surge during the respectiveSCR commutation starts of relatively large magnitude current pulses tothe respective motor windings. This entails higher ratings for thecommutation circuit components. SCR commutation is a problem therein,with miscommutation a serious factor. More than three times the energyis required for commutating their respective SCRs to off, than in normalSCR control circuits. Bifilar motor control systems generally requirepulse width modulation circuits for stabilization, correspondinglyincreasing the number of components.

SUMMARY OF THE INVENTION The polyphase motor/control systems of thepresent invention overcome the aforesaid limitations of the prior art.Associated windings pairs are arranged for each phase on a full pitchand consequent pole basis. The two windings of each pair are arrangedapart magnetically. Minimum coupling is provided between them. Currentsare supplied 180 apart electrically to the complementary windings ofeach phase pair producing a rotating polyphase field. Their SCRcommutation requires only normal size components and power, with simplercircuitry. The full pitch windings of the motor hereof carry current inopposing directions in any stator slot, resulting in no loss in windingeffectiveness or reduction in magnetic efficiency. The complementarywindings use common stator slots with 180 phase separation thereat. Therelatively low magnetic coupling between the associated windings of eachwinding pair per phase hereof is most advantageous in effectingpractical SCR commutation with inherent self-clearing for the SCRs eachcycle through energy from the motor. These windings are connected in amanner to perform magnetic inversion, with a zero average value magneticwave moving around the air gap. Such magnetic wave is similar to thatoccurring in three-phase squirrel cage motors.

The invention control system, as will be shown hereinafter, is far lesssusceptible to miscommutation than prior motor/control systems. Also, itis far less vulnerable to damage of its SCRs and control circuits shouldmiscommutation occur. If miscommutation should take place a motorwinding is in series with each SCR, and its inductance limits themaximum current that may be drawn through the SCRs to approximately onlythat of the motor stall current. Such nominal overload is a factorreadily designed for. The invention pulse and control circuitry is fullycoordinated with the motor winding reactions to provide the SCRself-clearing action, and is quite foolproof, as will be set forth. Thefull pitched consequent pole polyphase winding array of the motorshereof are not voltage limited in the stator slots. Full phaseinsulation is incorporated. This affords a significant breakthrough forlarge motor power ratings with the invention motor/control system.

The motor systems hereof are operated off a DC supply. Such supply isderived from a single-phase or a polyphase power source, that isrectified. At rated load and speed the motor/control systems present apower factor of 0.90 to the AC line where it is single-phase, and 0.95to a three-phase line. Such power factors are better than conventionalpolyphase motors present di- A rectly to the power line. Their overallefficiencies are comparable to three-phase induction motors undersimilar operating conditions. The motors are readily stopped by theassociate control circuitry. The system electronic circuitry isreliable, and less in cost than in comparable motor control systems.Different control circuits effect various predetermined operationalmodes for the invention motor system. The stator winding array isthoroughly integrated with the electronic pulse and control circuitry toprovide the novel, efficient and self-clearing induction motor systemhereof. Its wide flexibility in motor power rating, speed range, andoperational control establishes an important new branch in motortechnology.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a schematic circuit diagramof an exemplary embodiment of a three-phase induction motor andassociated electronic pulse and control circuitry.

FIG. 2A is a developed diagrammatic representation of the consequentfull pitch complementary windings of the exemplary three-phase inductionmotor represented in FIG. 1.

FIG. 2B is a cross-sectional view illustrative of the winding turns inslots of the exemplary stator per FIG. 2A.

FIG. 3 is a graph showing of the stepped voltage that appears across therespective windings of the stator.

FIG 4 is a graph showing the typical current flow through the power SCRsof the motor system of FIG. 1.

FIG. Sis an electrical circuit diagram of an integrated phase of thethree-phase motor/control system of FIG. I, used in explanation of theinvention system.-

FIGS. 6A and 6B illustrate the SCR triggering-on pattern of themotor/control system of FIGS. 1 and 4.

FIG. 7 is an elevational view of an induction motor and its control unitin accordance with the present invention.

FIG. 8 is an overall circuit diagram of a motor/control system hereof,with one form of variable dc voltage supply.

FIG. 9 is a family of operational motor torque/speed curves, as derivedwith the control system of FIG. 8.

FIG. 10 is a family of curves of typical motor systems hereof, over wideoperational ranges of torque and speed.

FIG. 11 is a family of curves of the constant horsepower operationalmode of the motor system hereof.

FIG. 12 is a circuit diagram of a control section that provides theconstant horsepower operation of the motor system per FIG. 11.

FIGS. I3 and 14 are families of operational curves of motor systemshereof with constant torque output characteristics.

FIG. 15 is a graph of the compensation applied by the control section ofthe motor systems that provide the constant torque output modes of FIGS.13 and 14.

FIG. 16 is an electrical circuit diagram of the control section for themotor/control system hereof that pro- FIG. "is a family of curves in thetraction operational mode of a motor/control system hereof.

FIG. 18 is a diagram used to explain the traction mode of operation ofthe motor system.

FIG. 19 is an overall schematic diagram of a motor/- control systemhereof that effects traction operational modes in accordance with FIGS.17 and I8.

I: Basic Motor and Control Circuitry The polyphase motor system issupplied from a unidirectional power source. Reference is made toschematic circuit FIG. 1 of an exemplary motor/control system. Thedirect current source voltage +V is applied through contactor I9 to lead20. An'interphase transformer 21 is used between the motor windings anddirect current lead 20, at its mid-point, to reduce third harmoniceffects. The motor windings, as will be set forth, are in complementarysets energized respectively through leads 22 and 23 from the interphasetransformer. The six windings of the exemplary three-phase stator,indicated at 24, are grouped through their terminals, as follows: N1, N3and N5 to lead 22; and N2, N4 and N6 to lead 23. Such dual connection ofterminals N1 to N6 requires only two external motor leads therefor. Theopposite terminals of the respective stator windings are Ll through L6which connect directly to corresponding terminals 1 through 6 of theelectronic SCR pulse circuitry, as indicated. The stator windings are insix sets 4), through 4),, each termed half-phase" herein. Their physicalconstruction and arrangement in stator 24 is described in more detailhereinafter in connectionwith FIGS. 2A and 2B.

The sets qS through (1),, are'wound as complementary pairs, eachhalf-phase pairproviding the electrical and magnetic resultant of onefull phase for the motor system. The successive unidirectional pulses tothe stator windings are of 180 duration electrically, and aresequentially applied in the same direction to all the windings. Thewindings of each half-phase pair are arranged 180 apart magneticallywith respect to each other. The preferred winding array is full pitchedand consequent pole for maximum efficiency and for effective foolproofoperation of the system. The six windings (I), through d), arecoordinated to effectively provide the three basic phases 4),, 4 for thethree-phase induction motor hereof.

Referring to half-phase windings (b, as reference for the magneticsequence of the windings in stator 24, its reference position isindicated as 25-1. The second winding 41', at position 25-2 is 60magnetically apart from the reference winding 41,. In like manner, thethird winding 41;, at its reference start position 25-3 is an additional60 apart from the second winding d), magnetically, and therefor apartfrom reference winding The fourth winding d), at its position 254 is 60'apart magnetically from its adjacent winding 3,

and correspondingly apart from reference winding (b Windings d), and 4:,are used as paired, complementary windings to constitute stator phase1b,, at 180 MMF separation as indicated at 26,27. The fifth winding cb,is 60 apart magnetically from the fourth winding d) and correspondingly240 MMF apart from reference winding da, It is noted that fifthwindingqlv, is 180.

magnetically apart from second winding and forms complementary windingpair phase 4), for the motor/- control system hereof; The sixth winding4: is set 300 magnetically apart from the reference winding do it being60 apart from the fifth winding Winding 4a, is 180 apart magneticallyfrom third winding 4a,, and with it fonns the third phase set du Anactual winding diagram of the aforesaid six windings in a stator isshown in FIG. 2A.

A power silicon controlled rectifier is connected in series with eachwinding half-phase (I), through 4: These SCRs have their respectivecathodes at ground line potential (33), with their anodes connectingdirectly to the respective stator windings. The windings d1, through 4;,in turn connect to the positive DC supply +V through interphasetransformer 21. .The anode of SCR 30-1 connects to stator winding 4),through terminals 1 and L1. Similarly, SCR 30-2 connects to phasewinding 4:, through-terminals 2 and L2. In like manner SCRs 30-3 through30-6 are connected in series with respective half-phase windings air,through 4:, between common ground 33 and the DC buss 20. As will beshown in connection with FIGS. 6A and 6B, the current pulses through theSCRs and the associated half-phase windings are initiated over 180electrical degree periods for each SCR/winding unit. Also, theseare-effected at 180 electrical separation and in the same direction inrespective complementary SCR phase windings. Such complementary 180electrically and 180 magnetically separated pairs are the 1 and 4 unitsthat comprise polyphase motor phase di the 2 and 5 units for phase bu;and the 3 and 6 units for phase 4),. It is noted that phase set qSutilizes half-phase winding which is displaced magnetically 240 from thereference phase (15,, and that phase set (b incorporates halfphasewinding (1);, at 120 relative to reference phase 4),. Thus 120separation between the three basic phases of the three-phase motor isprovided herein. By reversing the control sequences, as for phase (1),,and phase 11 in the pulse system, we reverse the direction of rotationof the motor per se.

Each SCR 30-1 through 30-6 has a reactive diode 31-1 through 31-6respectively connected across it. These diodes perform several importantfunctions in the operation of the system, as will be described. A simplecommutating circuit is connected between the two SCRs of eachcomplementary phase pair, such circuit comprising an inductor andcapacitor in series between the respective SCR anodes. The SCR pair30-1, 30-4 have commutating inductor 28-1 in series with commutatingcapacitor 29-1 thereat for phase 4) Similarly, inductor 28-2 in serieswith capacitor 29-2, is used in SCR pair 30-5, 30-2 of basic Commutatinginductor 28-3 and commutating capacitor 29-3 are connected across theanodes of (b phase SCRs 30-3, 30-6. The interaction of the commutationcircuits, the reactive diodes, the SCRs and the respective statorwindings are described in detail in connection with FIGS. 4 and 5.

An SCR gate driver and sequential triggering circuit (35) is utilized toprogram the operation of SCR's 30-1 through 30-6 hereof. Each gate ofthe-SCRs connects to SCR gate driver system 35 through respective leads32-1, through 32-6. The driver and logic circuit array 35 is energizedby low unidirectional voltage +V preferably regulated to maintainaccurate operation. The electronic clock 40 is similarly energized by +Vand is connected to the logic circuits 35 in a well known manner vialead 36 for their scr gating control. The exemplary clock is aunijunction transistor 47 as illustrated in FIG. 5. However, otherreference clock circuits may instead be used. The gate drive operationof the silicon control rectifiers hereof, in relation to the clockfrequency and the SCR gating is described in detail hereinafter inconnection with FIGS. 6A and 68.

Different performance modes for the motor of the invention system areaccomplished by the use of different control circuits for operating thebasic clock 40. The frequency of the clock 40 determines the rate ofsequential gating of the SCRs in a directly proportional way throughconventional logic circuits in unit 35. In the exemplary system of FIG.1 the uni-junction transistor within clock 40 (see FIG. 5) is controlledas follows: The operating DC voltage +V, is applied to diode 39 throughlead 20'. Connection point 41 between diode 39 and electrolyticcapacitor 42 extends to clock 40 through a control resistance chain,Exemplary capacitor 42 is rated at microfarads, with one side grounded.Its rating may be in the range of 2 to 30 microfarads. Resistance units43, 44, 45 are connected in series between point 41 and the uni-junctiontransistor (47) in clock 40(see FIG. 5).

In the exemplary circuit potentiometer 43 is 35,000 ohms; potentiometer44 is 50,000 ohms, and resistor 45 is 33,000 ohms. Thus less than 1 wattflows in this resis-, tance chain for the indicated volts for +V,,. Thefrequency of clock 40 is proportional to the applied voltage +V,,, andinversely proportional to the total of resistances 43, 44 and 45 inseries to the clock. A small capacitor 46 connects between resistor 45and ground, as 2 microfarads. The clock frequency is inverselyproportional to the capacitance of element 46. Capacitor 46 is preset atthe factory to remain constant. As indicated, the voltage bias +V forthe uni-junction clock 40 isregulated to remain substantially constantin the field. Thus, with the buss voltage set at +V,, as at 120 volts,control of the speed of the motor basically depends upon resistancechain 43, 44, 45. In practice, resistor 45 is fixed and potentiometer 44is held at its factory setting. Potentiometer 44 is a frequency trimmingunit. Its setting in conjunction with the value of the capacitor 46 andresistor 45 is preset to determine the minimum motor speed. Essentiallythen the speed of the motor is varied by controlling potentiometer 43.As the resistance setting of control potentiometer 43 is reduced themotor speed is increased. Increasing +V,, also increases motor speed.

Reduction of the net resistance in resistance chain 43, 44, 45 resultsin increase of the clock frequency, and in turn the rate of pulsing ofSCRs 30-1 through 30-6 at a corresponding rate. The rotating magneticfield created by the effective polyphase stator 24 is at a speed thatcorresponds to the number of magnetic poles and to the frequency appliedthrough the SCR control. The induction rotor within the stator rotatesat the corresponding speed as in induction motors. Accordingly, themotor system of FIG. 1 with a fixed +V, buss essentially has its speedcontrol through potentiometer 43, which in this mode acts similar to afield control in a direct current motor. The potentiometer 44 andresistor 45 are selected to present a resistance in the circuit to theunijunction transistor, which in conjunction with capacitor 46 presets asafe minimum operating speed for the motor hereof. Adjustment in thefactory of potentiometer 44 and of condenser 46 presets the control forparticular motors supplied with the control system. The controlparameters stated are operated at low voltage and very low power, andare effective for fractional to high horsepower motors.

The motor is started up by maintaining the control potentiometer 43 atits low" speed position, and closing contactor 19 to apply buss voltage+V, to the system. The clock 40 is thereby energized to in turn controlthe sequential triggering and logic circuits 35 for the gate driveraction of the SCRs 30-]! through 30-6. The control potentiometer 43 isthen moved to effect the operating RPM desired. When the motor is to beshut off, contactor 19 is opened to remove the buss voltage +V,,. Thepresence of electrolytic capacitor 42 prevents the clock frequency fromgoing immediately to zero and thereby stop the motor too fast. Instead,capacitor 42 permits the clock frequency to run down gradually. Duetothe prolonged discharge time of capacitor 42, it maintains a continuallydecreasing voltage to the uni-junction transistor in clock 40. It is animportant practical way to stop the motor. It is in a positive controlmode all the way, in lowering control fre- I quency directly to operatethe motor system correspondingly to slow then stop. There is thus noabrupt or dangerous system action possible due to any direct shut-off orloss of dc source power.

Another way to stop the motor is to gradually slow it down via controlpotentiometer 43 to a low speed, then remove the +V,, buss by openingcontactor 19. Further, the motor may be stopped through eddy currentbraking' by directly shorting-out the action of condenser 42 and theresistance chain. This is done by grounding the uni-junction transistor,as by closing switch 37 indicated in FIG. 5. This type of braking occurssince three of the SCRsremain on when the commutation is abruptedlywithdrawn. The other aforesaid motor stopping ways however are moreeffective than due to eddy currents flowing in the windings.

ll: Basic Motor Winding Arrangement FIG. 2A is a developmentdiagram ofthe six halfphase windings d), in the stator 24 of the exemplaryinduction motor. Each full phase set (1),, (b is comprised of twoindividual half-phase windings, as previously set forth. The windings ofphase 4),, are arranged to form the four magnetic poles (l), (2), (3),(4) indicated on the top of the diagram. Those of phases 4),, and (1),,are arranged to form respective four pole sets. The pole sets of thethree effective phases are arrayed at 120 magnetic separation forthree-phase operation. The stator has 36 slots as indicated. These slotsare arranged in a 360 cylindrical form, as is understood in the art. Therotor operates within the cylindrical stator opening. Half-phase sectionB, in the exemplary stator is formed as two concentric windings -1- and-7- in series between terminals N1 and L1. Winding -lextends from slot 1to 12. Its effective MMF poles (I) and (2), are indicated by the arrowsin slots 2 and 11, a spread of nine slots in the 36 slot stator; This is90 mechanically, which in the four-pole stator is 180 MMF poleseparation. Winding -lis effectively fully pitched, eventhough its turnsextend over adjacent slots.

The preferred windings for the stator hereof are full pitch for bestoperation and efficiency. However, somewhat less than full pitchwindings are also within the purview of the present invention. Thesecond winding portion of half-phase d), is concentric winding -7- inelectrical series with -l-. lts effective magnetic poles (3) and (4) liein stator slots 20 and 29, indicated by the arrows therein. It is notedthat the magnetic separation of MMF poles (2) and (3) also is 180, as aneffective mechanical separation of 90 exists between slots 1 l and 20.Correspondingly, effective MMF pole (4) in slot 29 is equal and oppositeto that in slot (2) of winding -l-, and is 180 MMF apart therefrom. Thestated magnetic pole separations is indicated in FIG. 2A adjacent to themagnetic pole arrows along the bottom of the do, windings. The currentdirection i indicated at terminal N1 is into winding -lwhen its SCR(30-1) is turned on. The positive DC supply flows into the N1 terminalin such 180 electrical period, as stated in connection with FIG. 1. Thewindings -land -7- are in series, and as illustrated extend over thestator in consequent pole array, i.e., the alternate winding, no-windingarray.

The half-phase action of windings dz, is due to their being utilizedelectrically and magnetically for only one-half cycle in the full phase1b,, operation. Its com-. panion half-phase windings -4- and -10- are inelecwinding set is separated 180 magnetically from its complementarywindings -land 7- of the (I), set. Thus the -4- concentric turnseffectively extend between MMF poles (2) and (3) at slots 11 and 20.Also, the second winding -10- of the 4: set extends between MMF poles(4) and (l), effectively at slots 29 and 2. It is further noted that thewinding progression and array of the set is the same as the d), setexcept for their 180 MMF separation at mechanically hereof. Also, theaway current pulse i, into terminal N4 provides the MMF effective poledirections as indicated by the heavy arrows in windings -4- and -10-.Their magnetic poles are in opposite relationship'to the adjacent MMFpole directions of the winding set -1- and 7-. Since the unidirectionalcurrent pulses into the 41 windings are 180 electrically behind thecurrent pulses into the d), windings, and flow into the windings in thesame direction, the effective magnetic pole array presented by thewinding pair and 41 provides the effective full phase (11,, action forthe'motor stator, as will now be understood by those skilled in the art.

Significantly, the consequent pole and interposed re lation of thecompanion windings of each full phase, as described, minimizes themutual induction between complementary windings. Their current pulsesare 180 apart electrically, and are directed to flow in oppositedirections in their respective turns in common slots, as denoted in FIG.2A by the resultant MMF arrows. Such arrangement further reducesmagnetic induction between the companion windings of each phase set. Thewinding turns at their end area, namely outside of the respective slots,are preferablyuncoupled to further minimize said mutual induction.System operation advantages accrue from the exemplary winding array withminimum mutual inductance between the windings of complementary sets d),and (b. of phase and likewise of phases 1b,, and (fa This is importantin effecting the cyclic self-clearing SCR commutation function herein.The inter-relationship between the energy of the rotating motor, theindividual stator windings hereof, and the SCR pulse circuitry providespositive SCR self-clearing, as described hereinafter in connection withFIGS. 4 and 5. More effective MMF results by making the windings as fullpitched as feasible, improving motor efficiency. The concentricallywound stator windings as illustrated may be replaced by lapp windings inpracticing the invention motor/control. Making the mutual inductionbetween complementary windings of each phase relatively low alsominimizes SCR commutation circuitry and power, as well as effecting theself-clearing and.

overall'foolproof SCR action by the present invention. This is inimportant contra-distinction over the prior art. Bifilar windings ofnecessity have close coupling of companion winding pairs whichmaximizestheir mutual induction, and create deleterious strong inductions to theSCR circuits.

The effective phase 4),, of the stator is provided by the 4;, and (bwinding sets. ,As indicated in FIG. 1 at 25-5 the half-phase d windingset -5- and -II- is 240 mag-.

netic degrees apart from that of reference windings (1),. The centralleft turn in winding is in slot 14 which is 12 slots beyond slot 2 forthe corresponding turn in the -1- reference winding hereof. This placesthe corresponding first magnetic pole for the -S- winding at 240magnetically apart from the (b, winding -lland its pole (1). The winding-ll1- is correspondingly 240 apart from the d), winding -7-. Thecompanion complementary windings -2- and -8- of half-phase set d1, is180 magnetically apart from the d), windings. These then form thecomplementary effective winding set for full phase (in, of the stator,in the same manner as hereinabove described for 4) In like manner the thphase is made up of the d), windings -3- and -9- that are 120magnetically apart from the first pole (1) of the 4, set. The windingsof (b are -6- and -12- which lie 180 magnetically apart from the 5,winding set.

It is noted that when in its conducting mode each half-phase winding set(it, through (b generates four MMF poles in successive alternatedirection, in the manner indicated by the arrows for sets qb, and 4: inFIG. 2A. Thus each half-phase winding directly provides four MMF poleseach 180 apart magnetically about the stator (24!). Should three-phaseAC power be applied to terminals L1, L3 and L5, with N1, N3 and N5interconnected as a neutral point, a full rotating magnetic field wouldbe generated in the stator, and with its rotor therein would operate asan induction motor. If instead, half-phase windings (#2, d), and (it,were connected in a similar manner, the motor would operate likewise.However, in the invention system, each winding set 4), through 4),, isenergized in successive half-cycles as aforesaid, with the complementaryhalf-phases operated alternately to provide the resultant full phasemotor action as phases 4),, (b

The winding array for the stator per FIG. 2A utilizes all 36 slots toform an effective compact winding pattern. The motor has four effectivepoles for each full effective phase (b da and 41,. The respective polesets are at 120 and 240 magnetically apart from the reference set (in,and ti), for the three-phase motor operation. Stators with more or lesspoles, in multiples of two, are similarly wound as will now beunderstood by those skilled in the art. Also, motors with other thanthree phases may likewise be constructed following the principles setout herein. More than single phase is needed to make the induction motorsystems hereof selfstarting. The stator is preferably constructed withan integral number of slots per pole per phase to provide for fullpitched windings. Where otherwise feasible, more than fewer slots aredesirable for improved winding distribution factor.

FIG. 2B is a cross-section of a portion of the stator 24 of the motorhereof. The stator is of quality steel laminations. The illustratedportion contains the slots, say and 11, of the array of FIG. 2A. Withinthese slots are turns of the windings -1- and 4-, indicated as W-1 andW-d. A wide range of turns may be used for the windings. For example,such windings each may range from 5 to 75 turns per slot, depending uponthe size, power, and operating characteristics of the desired motor.Their wire diameter corresponds to the current demands in the motoroperation, as is understood in the art. A significant advantage of thestator winding array of the present invention is that the turns of thecomplementary windings of each phase set 4),, (b and (b can be fullyphase insulated in their common slots. This is illustrated in FIG. 28 byinsulation 34 around the grouped winding turns W-1 and W4 in commonstator slots 10,111. Such full phase insulation means that motors ratedfor operation at 500 volts, and higher, can be practically constructed.The motors of the present invention are not voltage limited at theirstator windings, nor otherwise mechanically limited, and horsepowerratings to 500 horsepower, and higher, maybe built and operated hereof.

III: Operation of the Basic Motor/Control System FIG. 5 is a schematicelectrical diagram of full phase 4,, of the motor/control system inaccordance with FIG. 11. The respective half-phase winding sets (1), andd), are illustrated in their physical relationship corresponding tostator FIG. 2A. Half-phase 11 windings W-l and W-7 are connected inseries between line 22 at interphase transformer 21 and the anode of itsSCR 30-1. Similarly, half-phase 4, winding sets W4 and W-l0 in series isconnected between line 23 and SCR 30-4. Their respective MMF poles areat separation as marked by arrows at the windings at successivepositions a, b, c, and d. The d, and 45 windings are displaced by onepole set, namely by 180 MMF.

1f interphase transformer 21 were not used, source voltage +V, would beapplied to all the N terminals of the six half-phase windings. Suchmotor/control system would still be effective, but at reducedefficiency. The reason is that square wave voltage pulses would beapplied to the half-phase windings, with triplicate harmonicscirculating via the reactive diodes (31) to phases accepting power. This,would increase the internal 1 R and diode losses. The interphasetransformer 21 eliminates the third and related higher harmonics fromthe half-phase windings d), to d), by providing the main potential insix-step waveform across the respective half-phase winding sets. Themore advantageous stepped waveform is illustrated in FIG. 3.

Transformer 21 minimizes all zero sequence harmonics including thetriplicate harmonics, resulting in closer approximation of thefundamental or base frequency applied to the stator windings. Thesewindings function in concert as an autotransformer, and provide avoltage magnitude, from the center tap of transformer 21 across thehalf-phase windings to their respective L terminal at the SCRs, of twice+V The actual magnitude of the six-step wavefonn across these windingsper se, between their N and 1. terminals, is somewhat less. Transformer21 is a balancing reactor which allows the two common neutral points,represented by respective leads 22 and 23 of the connected statorwindings, at N, to float.

The reactive diodes 31 are in shunt across the SCR current switches 30.These diodes prevent unusually high voltages from occurring because ofsudden current turn off by the commutation. Such voltages could beproduced as the result of high L di/dt. The reactive current flowgenerated between windings d), and (b, for phase (b, due to switch-offof SCR 30-1 actually takes place through the companion reactive diode31-4, eliminating such problem in the circuit; and similarly for du and(1a,.

In order that the SCR promptly turn off in accordance with itsprogrammed pulsing, conventional commutation components are used, namelyinductor 28-1 and capacitor 29d between the complementary l and 4circuits. These serve to back bias a conducting SCR for a finiteduration of the order of several microseconds. Such period of backbiasing results from the natural ring frequency of the selected L-Ccommutating components 28-1, 29-1, as is well known. The size ofcapacitor 29-1 is dependent upon the magnitude of the current fiowinginsay the 30-1 SCR prior to and during the commutation. The magnitude ofsuch current in turn depends on the per phase impedances of the motorwinding sets 100 etc, and the amount of magnetic coupling through thestator between complementary half-phase windings, as (p, and d), in theH0. example.

When a motor has its stator windings quite closely coupled, as inbifilar motors, their inductances substantially cancel out. In suchmotor system the magnitude of the current through the SCRs would berelatively very high, as it would be limited substantially only by thelow resistances of the stator windings. This makes SCR commutationdifficult and unreliable as expressed hereinabove. However, the windingsof the respective half-phases d), and for full phase are in consequentpole array and intentionally arranged for minimum mutual inductancetherebetween. This allows the separate half-phase windings to maintaintheir separate inductances and rotor linkages. There results relativelyfast commutation action of their SCRs 30-1, 30-4. The SCR currents,power rating, and associate circuitry are thereby substantially reduced.Across each SCR is resistance-ca'pacitance network: 51-1, 52-1 acrossSCR 30-1, 51-4, 52-4 across SCR 30-4; etc. These R-C circuits aresnubbers, used to bypass spurious harmonic frequencies and stabilize theSCR control circuitry.

The SCR gate driver circuit 35 contain the sequential triggering logiccircuits for programming the SCR conduction periods in known manner.Individual control leads extend from circuit 35 through respectiveseries resistors as 54-1 and 54-4 shown in FIG. 5. A layer of gate driveamplifiers 55-1, 54-4 etc is used. These may be considered bufferamplifiers and/or amplifiers for gate drives of higher power SCRs. Thetransistors 55-1, 55-4 are biased by regulated constant voltage +V,,,,at line 58. The exemplary magnitude of +V,,,, is 5.0 volts, also usedfor the uni-junction transistor clock circuit 40. The coupling resistors54 connect to the base of the gate drive transistors 55, which in turnconnect to the gate electrodes of the associated SCRs 30 through leads56-1, 56-4. Loading resistors 57-1 and 574 are conventionally connectedin this amplifier circuit.

SCR circuits are subject to occassional turn-on at wrong times. Reasonsare varied, such as high line transients, an improper logic signal, etc.When this happens the commutating capacitor 29 may not be charged, andcommutation cannot occur in the normal manner. If for example, SCR 30-1and 30-4 were thus on" at the same time, they would each be in seriesconduction with their respective half-phase windings 4:, and cb, at thesame time. As these windings have low mutual inductance due to themanner of their winding and array as hereinabove set forth, the currentthrough their respective SCRs becomes only nominally high. In fact, thetotal current through the windings and the SCRs during suchmiscommutation does not exceed the stall torque current for the motor.The rating of the SCRs used are selected accordingly, a practicalbenefit. In control circuits, as inverters, where companion SCRs areconnected line-to-line to the DC source, miscommutation would result inpermanent damage to the SCRs. Often rather complex protective circuitsare provided therefor.

In the motor system hereof the low order of mutual coupling betweencomplementary half-phase windings allows normal commutation in one phaseto force commutate in the adjacent phase. Also, the reactive energy fromthe operating motor windings is discharged cyclically during eachrevolution to provide strong back bias for all the SCRs, as will now beset forth. Referring to FIGS. 1 and 5, let us assume that the SCRs ofsections 1, 5 and 3 are turned on and conducting in normal manner, innormal phase relation. Then in sequence SCR 30-4 of section 4 is turnedon." This normally would cause its companion SCR 30-1 to be commutatedto off." However, if 30-1 does not commutate to off it is in a latchedcondition, termed miscommutation. When such condition occurs thecommutating capacitor 29-1 between sections 1 and 4 has no storedenergy, and further normal commutation thereby in the companion sections1 and 4 cannot occur.

The logic circuit 35 driving the gates of SCRs 30-1 through 30-6 isignorant of this miscommutation, and proceeds to gate the section 2 SCR30-2 to on." As SCR 30-2 begins to conduct, the forced commutation turnsoff" companion SCR 30-5, and allows the forward current in half-phasewinding dz, to increase after such commutation between SCR 30-5 and30-2. However, before 'the forward current starts to rise in SCR 30-2,stored energy trapped in half-phase winding #1,, flows through reactivediode 31-2. Since sections 1 and 4 are short circuited across their SCRs30-1, 304 due to the stated miscommutation, the reactance of all theother windings linking the do, and d, windings through the stator isgreatly reduced from the normal state. During such substantially reducedreactance state in the motor, the reactive current that flows isrelatively high in amplitude and of short duration, of the order of 200milliseconds. The net effect of this strong reactive current flow is toback bias all the SCRs in the motor circuit, and return energy to the DCsource +V,,. As

. this energy is being returned to the source, all the power SCRs 30-1,30-4 etc are presented with a relatively long tum-off time. In effectenergy stored in the operating motor, magnetically and mechanically isutilized for this back biasing of all the SCRs in the next full cycle.During this turn-off or back biasing period all the conducting SCRs stoptheir conduction and go into a forward blocking voltage condition. Aftersaid reactive current flow, and the forward voltage is reapplied, themiscommutated or latched on SCR, namely 30-4 in this example, does notreturn to the conducting state since by then the sequential logicgatesignal therefor is at zero.

The sequence of events herein described provides automatic self-clearingof the SCRs before the following cyclic period of any latched on pair isover. lt is noted that any one section including an SCR 30 and itshalfphase winding set is in effective series relation with any othersection. The said reactive energy enters the on sections in the reversedirection which is just right for the self-clearing of the SCRs to offthat results. Thus, miscommutation directly brings on the strongreactive current flow and promptly clears it away. Also, there is adifference between such mode of SCR clearing and the normal mode ofcommutation as to the relative magnitude of the current flowing. Theimportant factor herein is that the six half-phase windings in thestator arrayed for effective three-phase operation has all the windingsinterlinked at the fundamental frequency across the stator with nowinding independent of the other. Also, the direction of the strongreactive inductions into the arrayed half-phase windings d), to 11),,hereof is such as to induce high voltages into the conducting SCRsections that safely and directly back bias them to off. For thisreason, when looking at the SCR forward voltages there is a negativegoing portion during each period of conduction for each phase, whichconducts reactive energy during a given cycle, resulting in the SCRcyclic conduction per FIG. 4 with a gap at g. The first turn-offpossibility is the normal programmed impulsed commutation initiated bythe SCR gate driver 35 with the commutation components 28,29 acrosscompanion SCR pairs. The second turnoff possibility is presented whenthe usual reactive energy hereof flows in that phase pair, indicated ate in FIG. 4. This does not occur if they are latched up. Two furtherturn-off possibilities for that particular condition occurs when theother associated phase pairs are commutated. A latched on pair iscleared by the strong reactive surge generated by the other windings asherein set forth. Overall, the combined commutation factors inherent inand compositely functioning'in the motor/control system 'hereof providespractical, effcient and substantially foolproof operation including itsinternal self-clearing commutation.

Reference is made to FIG. 4 which illustrates a typical SCR 30conduction pattern during its normal 180 electrical on" period in themotor/control system of the present invention. The initial commutatinggated pulse to the SCR programmed from the logic circuits 35 results inthe sharp narrow pulse a. The exemplary pulse a peaks near amperes and[has about a 2 duration in the 180 electrical conduction or on periodfor the particular SCR. The normal operating cyclic reactive energydischarge by the motor windings generates a sufficiently strongback-bias energy into this SCR section that is set for conductionthrough its half-phase windings (11 so as to directly create the coniduction gap g. Such gap 3 extends over the range of 15 to in the period.The SCR conduction thereupon resumes and rises slowly until about the140 point indicated at c. It peaks here at about 6 amperes, then droopsoff along d to about the 3 ampere level, when it is extinguished bynormal commutation at the 180 period end.

The triangular shaped conduction curve b, d is the combination of thereal current components in the associated winding (a while otherhalf-phases are turned on." Any one SCR section is not independent, asthe windings of the stator 24 are all linked magnetically across thestator structure, and the sharing of the load by the SCR sections thatare on. In normal operation three SCR sections are on" at any one time.Importantly, should any one SCR not turn off but instead latch on" asindicated by continued conduction at e beyond the 180, it is normallyherein directly extinguished in time by the 190 point. The reason isthat the same periodic back biasing effect in the system hereof thatproduces the SCR conduction gap g operates to back bias the latched on"phase at e. Should this not be thus effected in some instance, theoverall strong reactive energy back surge described overcomes it and isthereby self-cleared.

FIGS. 6A and 6B illustrate the logical gating action on the successiveSCRs 30-1 to 30-6 initiated under the control of the logic circuitry 35.It is understood that the basic or fundamental frequency that effect SCRperiods is derived from the frequency of the pulses that are generatedby clock 40. The clock frequency in turn is determined by controlcircuitry as 50, in the manner heretofore described. At a given clocksetting the gating pulses llT, 2T, etc are successively generated andimpressed in the time sequence, at 60 spacing herein, to the SCR gates.In FIG. 6A the regular pattern of the conduction periods for the SCRsections 1 through 6 are outlined. They are successively started at 60intervals in time, reckoned over 360 of the fundamental frequency. Theirduration is l80, when the SCR commutation occurs due to the activationof the complementary SCR section. The companion SCR sections are showngrouped together in FIG. 68; I and 4, 3 and 6, 5 and 2 respectivelyforming full phases 4: c, 4) b as aforesaid. As the SCR sections 4, 6and 2are conducting at 180 being their associated sections 1, 3 and 5(,per FIG. 6A), they are indicated in their equivalent negativedirection in FIG. 68. It is to be understood that FIGS. 6A and 6B areexplanatory timing pattern diagrams and that the actual SCR conductionflow corresponds to FIG. 4. I

The actual duration of each conduction period 3T is directlyproportional to the actual frequency of clock 40. The clock frequency isderived by the voltage and impedance applied-to the emitter of itsuni-junction transistor 47. An emitter follower amplifier 49 isoptionally used for transistor 47, with output at lead 36 and fixedbiasing of both through +V,,,, and coupling resistor 48. The comparableuni-junction transistor clock circuits of FIGS. 12 and 16 do not utilizean amplifier. Further, equivalent other types of clock circuits may beused. These control the logic circuits which include flip-flop circuitsof known configuration. As described in connection with FIG. 1, resistor45 is fixed and capacitor 46 is preset. Thecontrol unit 50 containspotentiometers 43, 44 and connects to the source voltage +V,, via lead20". Control knob 43' operates a contained speed control potentiometer43 for the motor operational purposes set forth hereinabove. The motoroperation with such control section will thereupon depend upon thesetting of 43' and the magnitude of +V,, applied.

FIG. 7 is an elevational view ofa motor/control in accordance with thepresent invention. The motor contains the squirrel cage type of rotorindicated at 5% rotatably mounted with output shaft 61. The statorcontains the six half-phase winding arrays qS as described in connectionwith FIGS. 2A and 2B. The motor housing is supported on base 62. The sixstator windings hereof are terminated in eight-wire cable 63 atconnection box 64. The control circuitry including the six SCRs 3f),-the reactive diodes 31, logic circuits 35, clock 40 etc are incorporatedin panel box 65, as are the rectifier and DC output voltage controller.Cable 66 is to the three-phase power line. The control knob 67corresponds to 43' in FIG. 5 to directly control the operating speed ofthe rotor 59, as will now be understood by those skilled in the art. Therotor is preferably a squirrel cage type. However, reluctance type, or

wound rotor induction motors may instead be used if indicated.

IV: Motor Output Characteristics: In General former, variac 70, isinterposed between a three-phase power line at effective voltage E,, andthree-phase rectifier 73. The setting of variac 70 is externallycontrolled through knob 71. Its three-phase output leads 72 are at thesame frequency as the power line, impressing balanced three-phasevariable voltage upon rectifier 73. The dc output of rectifier 73 isshunted by filtering condenser 74. Where required, a filter choke coil(not shown) is inserted in series in its output. The magnitude ofrectifiedDC voltage at output lead 75 corresponds to the setting of arm71 of variac 70.

Lead 75 is connected to centertap 76 of interphase transformer 77, theouter terminals of which connect to the two groups of stator windingterminals N in motor 80 via leads 78, 79. The stator of motor 80contains an array of half-phase winding sets corresponding to thosedescribed in connection with FIG. 2A hereinabove. With two complementaryhalf-phase winding sets per basic phase, six are used for thethree-phase hereof. The SCR circuits in unit 82 for the motor controlare directly connected to the half-phase windings by the schematicallyindicated cable 84. In turn, the

logic circuits 83 are similarly coupled to unit 82 for control of theSCR and electronic components, in the manner already described inconnection with FIGS. 1 and 5. The clock 85 contains uni-junctiontransistor 87 coupled to follower amplifier 88, and biased by the lowlevel constant DC voltage +V,,. Resistors 89 and 90 complete theamplifier circuit,'the output of which is' connected to the logiccircuits 83 by lead 86. i

The control section that determines the frequency of uni-junctiontransistor 87 comprises fixed resistor 91, variable resistor 92, andpreset capacitor 94. The DC system voltage +V is applied via lead 75 tothis section at resistor 91. Lead 93 couples this series control sectionto the emitter of uni-junction transistor 87, in the manner of thecorresponding sections in FIGS. 1 and 5. Variation of the resistancesetting of potentiometer 92 controls the frequency of clock 85; a higherresistance setting resulting in lower frequency; a lower resistancesetting, in higher frequency, as set forth hereinabove. Further, theuni-junction frequency herein is slaved to the voltage +V as set byvariac 70 so that at any given position of potentiometer 92 the.effective DC value (+V) impressed upon this control section changes thefrequency of the uni-junction transistor 87 correspondingly as well. Theclock 85 frequency is thus determined by the voltage setting per 71 aswell as the resistance setting at 92. The range of frequency controlherein, as provided by potentiometer 92 is often narrow, as a vernierbehind 71. For a particular setting of potentiometer 92 the clockfrequency 85 is thereby slaved to the magnitude of the voltage 0+Vsupplied to the motor 80. A higher voltage setting +V results in higherpower and torque for motor 80, as well as higher base frequencyoperation of the motor via SCR unit 82, in turn controlled by the logic83 and clock 85 units. Conversely, a lower voltage setting +V produceslower effective torque and power output by the motor, which is thus alsooperated at a lower base frequency via clock 85. The application of suchcoordinated control 71, 92 on the motor hereof is now described inconnection with FIG. 9. i

The motor 80 of the system FIG. 8 typically produces the family ofcurves illustrated in FIG. 9. Basically, the torque axis is a functionof the input DC buss volt-age +V and the frequency of the clock 85..Itshorizontal speed axis n is a function of the clock frequency only. Bythe arrangement hereof that keeps the clock frequency proportional tochange in input DC buss voltage, there results the family oftorque-speed curves of FIG. 9. With potentiometer 82 present, thedifferent +V voltages applied result in the characteristic spacedcurves, herein at 20 volts apart. The top 120V curve is highest of thisgroup, and produces the highest torque, namely 7 pounds-feet by a onehorsepower motor. The rated I-IP output occurs at about 1,700 RPM, asindicated. By varying the motor speed via potentiometer 92, with +Vpreset at 120 volts, the rated I-IP output occurs over the narrow rangealong the broken lines indicated, namely from 1,400 to 1,900 RPM. Ratedhorsepower prevails in this continuous range of speeds, at the downwardbend of the torque-speed curve. This practical operating range of speedsfor the 120V DC characteristic is controlled in vernier fashion over therated indicated horsepower range, through settings on potentiometer 92,as will now be understood.

Corresponding operating ranges for any voltage setting +V are set forcontinuous motor operation. Ope ration at speeds other than in suchrange is limited to relatively short periods to prevent excessiveheating in the motor. Continuous operation at somewhat higher speeds isfeasible. However, due to the downward slope of the torque-speed curve,the operating speed range is limited for a given voltage +V, as stated.The family of curves of FIG. 9 is developed with operational DC bussvoltages asindicated, in 20 volt steps down to 20 volts DC. It is notedthat lower nominal torque levels occur at corresponding lower operatingdc voltages +V. Each of such curves nevertheless has similartorque-speed operational characteristics, at correspondingly lowerhorsepower potential as buss voltage +V is lower. The dashed lined curveA in FIG. 9 is the operating locus for varying buss-voltage +V, withcorresponding clock frequency slaved thereto (per FIG. 8). The curve Ais generated with potentiometer 92 setting held fixed. The variac 70 isadjusted for voltage settings, through arm 71. The resultant A curveshows, load output and torque increasing with motor speed n, both withvoltage +V. The motor may operate continuously at any point under the Acurve over the entire speed range.

The overall system performance of the motor/control hereof is a functionof the electromagnetic design of the motor in a manner in which itsspeed is regulated by the control section, as set forth. Electromagneticdesign of the motor determines the shape of the fundamental torque-speedcurve for a particular motor. FIG. 10 illustrates the characteristicmotor torque-speed curves of various motors, and at several quadrants ofoperation. The first quadrant contains the family of five curves a, b,c, d and e, all in positive torque positive speed relation. In analyzingthe quadrant (1.).torquespeed characteristic, the motor system hereofmay be compared to a conventional three-phase induction motor insofar asits inherent torque-speed curve is concerned for a particular voltageand particular frequency applied. The'different curves a through 2 showthe effect of differing resistanceof the induction motors, inconventional as well as in motor systems hereof. The induction typemotor with squirrel cage rotor has a fixed effective resistance valueset by its design and construction. Conventional AC induction motors foroperation off polyphase power lines, are often designed to operate percurve b since this represents a compromise between good starting torqueand low rotor operating losses. Curve a for example represents a motorwith lower starting torque but also with lower rotor losses at fullload. Sustained operation of conventional constant-voltageconstant-frequency polyphase motors is feasible with slip, thus belowbut near what would be synchronous speed. When such motors try speed ofoperation at substantially less, the motor overheats and becomesdamaged.

The brushless DC operated motors of the present invention have the sameinherent torque-speed curves as a through 2 of FIG. l but with a largeadvantage over conventional motors off a power line. Both the speed andthe torque of the FIG. curves for the motors hereof are widelyadjustable instead of being fixed (including at negative values). Ashereinabove set forth, the torque is a function of the input DC bussvoltage +V and the accompanying slaved clock frequency 85; the speed n,a direct function of the clock frequency along. By maintaining the clockfrequency proportional to the input dc voltage +V per FIG. 8, thetorque-speed curves of FIG. 9 result, as hereinabove described. The highstarting torque curves of FIG. 9 correspond to curve 0 of FIG. 10. Theother curves in quadrant (l) have substantiallylower starting torques,or torques that are unsustained. Utilizing the control sequence per thecontrol section of FIG. 8, the first quadrant forward motoring curve isrepresented by the family of curves for FIG. 9. The particular curve ofcourse depends on the rating of the motor and the level of the operatingdc buss voltage +V applied, as will now be understood by those skilledin the art. In FIG. 10 nominal peak torque levels, and rated oroperating speed are indicated.

Should the positive motor rotation exceed the nominal synchronous speedN, value, it would be operating in the second quadrant (2). Theoperation of the motor would be in the regenerative or dynamic brakingmode. Such over speed produces negative torque as quadrant (2) presents,and involves speeds up to twice the inherent synchronous speed N Suchovershooting of the motor speed generates negative torque by the motor.The third quadrant (3) with both reverse rotation n, and negativetorque, is the equivalent of reverse motoring or generator action by themotor system. This mode of operation returns dc energy to the source 73.In the fourth quadrant, with reverse rotation n and positive torque, wehave the plugging operational mode. The motor/control systems hereof areoperable in all of the four torque-speed quadrants indicated in FIG. 10with controls and currents capability in reverse. Any particularoperating motor curve utilized in these quadrants, insofar as theircharacteristic curve shape and operation is concerned, depends upon therelative rotor resistance of the motor.

The flexibility of the motor system hereof as to performance andoperational modes is wide and varied. Infinitely variable speed controlis practical in course or vernier action, as set forth. The operation ofthe motors is relatively foolproof, with self-clearing commutatingaction. Rotating fields are generatedin the poly- K8 phase manner in thedirection that determines the rotor rotation in the (ll) quadrant.Reversal of the motor is readily. accomplished by the low power lowvoltage control section by simply reversing the logic action on therelative phases d (I), and (1),. Other important torque-speed operatingmodes than are described for FIGS. 8 to 10 are practical and feasible,together with their comparable action in the four torque-speedquadrants. In all modes it is preferred to utilize the high startingtorque that the invention motors inherently have. Also, they havesignificantly high stall torques without damaging the electroniccircuits, as aforesaid. In practice, the overall cost at greater than 2HP ratings has been found less than that of comparable DC motors.Besides, the motor systems hereof are explosion proof, and can bereadily hermetically sealed for indicated installations. Their size andweight per output horsepower, and overall system efficiency arecomparable. to three-phase induction motors at the same rated operatingload and speed. The operating speeds hereof are readily controllable andvariable, and unrelated to the ac line frequency which is rectified.Design speeds depend mainly upon the mechanical aspects of the motorconstruction, mainly its rotor stress and bearing life. Speeds of over40,000 RPM are practical. Further, the motor is not voltage limited inits construction, and systems with horsepower ratings above 500 arepractical.

V: Constant Horsepower Operational Mode FIG. ll containsthe family ofcurves A, B, and C with output torque vs shaft speed n in the constanthorsepower output mode. FIG. 12 illustrates the control circuit for themotor system hereof for deriving such mode. To provide for such constanthorsepower operation, the buss voltage +V,, to the motor system, forexample that of FIGS. I and 8, is held at a constant magnitude duringthis motor operation mode. The buss voltage +V,, in the exemplarycircuit FIG. 12 is therefor not connected to the control sectionthereof. The resistance chain 105,106 of the clock I00 circuit controlconnects to the fixed biasing voltage +V,,,,. The resistance ofpotentiometer 06 in this circuit is adjustable, to control the frequencyof uni-junction transistor 101, which in turn is transmitted to thelogic circuit via lead 108. The operating resistors I02, H03 are in thetransistor circuit between +V,,,, and ground, with output lead 108across resistor T03. The preset capacitor 107 connects with the emitterof uni-junction transistor I01 and the controlresistance chain 105,106and to ground potential. Capacitor I07 is preset as in the prior controlsections.

For a given voltage setting of the buss +V,,, say at volts, the familyof curves a, b, c, d result at specific motor speeds it set by speedcontrol 106. The peak torques of this curve family are the nominaltorque, and represent an envelope higher to and parallel to theoperating A curve. Curve A is for the said 120 volt setting of +V,,; thepractical operation at this voltage over its speed range n. For thismode of operation there is a minimum permissible or base speed asdetermined by the flux limit for the steel built into the motor. Themotor speed is not dependent on the selected buss voltage +V,,, butrather on the clock E00 frequency. The operating curves A, B, Crepresent constant horsepower output of the motor shaft at selected bussvoltage. At lower buss voltage input +V,, to the motor, a lower torqueoperating curve results. Constant horsepower curve B represents forexample a buss voltage +V,, of 60 volts impressed upon the motor system,with the +V,, biasing voltage kept constant, as at 5.0 volts. The motorspeed is effected via potentiometer 106, as will nowbe understood.

It is to be understood that the constant horsepower outputcharacteristic operation of the motor, as denoted by FIG. 11, resultsfor given buss voltages +V,, maintained on the motor with the motorspeed controlled at the uni-junction transistor or clock frequencycontrol, potentiometer 106 herein. Equivalent circuit arrangements forthis purpose may of course be used. The result denoted by the operatingcurves A, B and C for the selected buss voltages is similar to effectingfield control on conventional dc motors. The constant horsepoweroperating range presented by these curves are limited to the speedsindicated by their curves A, B, C. These all have minimum or base speedsbelow which they should not be operated for any considerable time. Themotor may be operated continuously under any combination of speed andtorque represented by the area underneath a respective curve A, B or C.

The reason the motor hereof when operated in accordance with FIGS. 1-1and 12 develops the locus of peak torque along a constant horsepowerline is because the motor impedance at peak torque remains essentiallyconstant, slip frequency being self-compensating. This factor combinedwith the factor that flux is inversely proportional to frequency appliedto the motor, causes the torque to run inversely proportional tofrequency, while the speed is proportional to frequency. This is becausethe torque is essentially a product of flux density times current of themotor. Thus it is seen that the product to torque and motor RPM is aconstant, and hence the resultant horsepower operation hereof is at theconstant mode.

VI: Constant Torque Operational Mode Reference is made to FIGS. 13through 16. These figures illustrate the operation and the controlsection for providing constant output torque with the motor systemhereof, at variable speed. FIGS. 13 and 14 illustrate respective ratedoperating constant torque output curves A and B of motors havingdifferent rotor resistance. The latter accounts for the different shapeof their torque-speed curves. The output curves A and B are similar tothose obtained by armature control on conventional DC motors, effectingconstant torque output throughout the speed rangerlt is noted 'thatthese curves actually provide rated torque at the output shaft all theway down to zero RPM. The family of curves a through e of FIG. 13correspond to rotor impedance providing output torque-speedcharacteristics per curve c in quadrant (1) OF FIG. 10. Thecorresponding curves a through e' of FIG. 14 relate to rotor resistanceproviding curve b of FIG. 10. The respective peak torques are at f andf' The rated torque curves A and B are at substantially less torquelevels for continuous operation. The downward sloped leg BB of torquecurve B is an expanded indication of the torque curve B when extendedbeyond the basic synchronous speed To provide the constant torque outputoperation per FIGS. 13 and 14, compensation is provided in the controlsection, per FIG. 16. Such compensation corresponds to the product ofmotor current and its impedance IZ. This is used to overcome theresistive component of the motor which is fixed while the buss voltage+V is varied. To maintain constant torque output, the product(V,,,/f)-I,, is held constant. The fundamental motor frequency is f. Thecurrent that the motor draws is I and V,,, is the voltage for the motorwhen its internal voltage drop is subtracted from the +V applied. Themotor frequency (f) resultant in accordance with the compensationindicated per curve C in FIG. 15, provides the constant torque outputcharacteristic A and B.'Such frequency compensation curve is effected bythe control section of FIG. 16 in conjunction with the varying appliedbuss voltage +V. Such result is obtained by placing Zener diode 117 inthe circuit of clock 110. It allows the clock frequency, namely that ofits uni-junction transistor 111 to change its frequency in a linearmanner C above dashed level D vs applied buss voltage magnitude +V. Thereference line D corresponds to the internal motor IZ voltage dropreferred to. The offset a corresponds to the slip required for maximumtorque.

The control section FIG. 16 applies the resultant frequency fromuni-junction transistor 111 via output lead 114 to the logic circuit.The resistors 1l2 and-113 complete the uni-junction transistor operatingcircuit, duly biased by fixed low voltage +V,,,,. A potentiometer 115 isin the input circuit to the emitter of transistor 111, and connects withvariable buss voltage +V. The variable +V potential may be derived froman arrangement per variac and three-phase rectifier 73 of FIG. 8, ascorrespondingly provided in FIG. 19, or otherwise. The setting ofpotentiometer 115 is unchanged when constant torque output is desired.On the other hand, the horsepower rating of the output can be controlledthrough the setting of potentiometer 11 5 to superimpose constanthorsepower control in conjunction with any torque setting. The Zennerdiode 117 is in series with potentiometer 115 and fixed resistor 118, tothe emitter of transistor. 11]. The preset condenser 119 connects theemitter to ground. A capacitor 1'20 connects between the Zener diode atlead 116 and ground, and is adjusted to provide desired rate of themotor system. It is noted that the frequency output of clock is slavedto voltage +V applied to the system, with the practical linear relationC over the operational speeds, but effectively lower due to the level Destablished by the Zener diode 117 therefor.

The IZ component includes the internal effective motor resistanceperconventional equivalent circuit analysis. If the motor had noequivalent resistance at all, constant torque output would occur bysimply vary- 1 ing the applied buss voltage +V, without the Zener diode.This is because the synchronous frequency of the motor would beproportional to +V, as the clock frequency would be derived directlyfrom such voltage input. However, as explained above, the compensated Cmotor frequency is used to overcome the internal IZ drop to effect theconstant torque output characteristics per A and B of FIGS. 14, 15. TheZener diode 117 in series with the resistance chain 115, 118 to theemitter of the uni-junction transistor 111 does not function as avoltage regulator. Instead, the Zener diode in the clock control circuithereof causes the uni-junction transistor, and therefor the clockfrequency output, to function as though the applied buss voltage +V werelower by a predetermined amount. The compensated frequency resultantaccording to curve C of FIG. 15 is provided by the exemplary controlsection including the series Zener diode 117 in clock 110 to accomplishthe constant torque mode.

VII: Traction Operational Mode Reference is made to FIGS. 17, I8 and 19in presenting the traction mode for the motor system hereof. Usualtraction motors are direct current machines with field and armatureconnected in series. As the armature speeds up its back EMF increases,that decreases current and weakens the field. As a result of theweakened field strength, the armature speeds up, increasing its back EMFto the level of the input voltage. This in turn weakens the field again.This process gives rise to the torque-speed characteristic of a seriesor traction DC motor. The invention motor system can advantageously becontrolled to simulate such traction characteristics. FIG. 17illustrates in curves A, B and C such mode, as derived from the systemschematically illustrated in FIG. 19.

The motor systems hereof produce very high starting torques, withsquirrel cage induction type rotors. Control sections are utilized thatprovide typically traction torque-speed type of curves as A, B andC. Asindicated, higher traction curve A corresponds to constant DC appliedvoltage of 140 volts for +V; next curve B, by 120 volts; and lower curveC by 60 volts dc. It is understood that other applied buss voltages +Vwill produce corresponding traction curves in the family per FIG. 17.Advantageously, unlike series-wound DC motors, the traction motor systemhereof cannot run away under no-load conditions. Also, since theirsynchronous speeds are controlled electronically, the motors hereof arereadily speed limited without circuit breakers. This affords asubstantial-safety advantage over series-wound DC motors, and alsopermits the traction drives hereof to be used in applications thatrequire occasional operation at even no load. By varying the DC bussvoltage to the traction motor system of FIG. 19 in a manner to bedescribed, a range of traction mode curves as A, B and C results. Thedashed line curve T are positions below the corresponding curves A, B, Cthe motor may be safely used in continuous operation.

FIG. 19 is a schematic electrical diagram of an exemplary motor/controlsystem hereof including control section therefor for operation in thetraction mode. The basic DC buss voltage +V is provided by threephasebridge with SCR voltage control unit 125 energized by the three-phasepower line at voltage E The output voltage of three-phase rectifier 125is phasecontrolled in conventional manner by three-phase SCR firingcircuit 126. A variable resistor 127 of circuit 126 is used to controlits operation on bridge 1 25, and in turn controls the magnitude ofunidirectional output dc buss voltage +V. A filter choke coil 1218 andshunt filter capacitor 12? is utilized with advantage at the outputrectifier 125. Buss DC voltage +V connects to the center tap ofinterphase transformer I30, the outer terminals of which are connectedby leads I31, 132 to the two floating neautrals N of the stator windingswithin motor 135. A squirrel cage type rotor is indicated at 134 withinmotor I35. The basic SClR motor control unit 136 connects to thehalf-phase windings at terminal board I33 of the motor via cable I37.Correspondingly the logic control circuits 138 are connected by cableI39 to SCR motor control unit 136. The unijunction transistor clock 140is coupled to control logic unit 138 via lead MI. It is understood thatSCR control 136, logic unit 138 and clock I40 have circuits andarrangements corresponding to those described herein- 1 above inconnection with FIGS. B. and 5. The bias voltage +V regulated to asubstantially constant magnitude, connects via lead M2 to theappropriate connection in the clock via lead 143, and in the logic unitvia lead ll l -tl.

A tachometer M5 mounted on shaft I46 of motor I35, accordingly rotatesat a speed equal to or corresponding to that of the motor. Its outputmay be direct current, or rectified AC, at voltage e applied betweenground M7 and lead M8 to terminal I419. The control voltage +e varies inaccordance with the motor speed, and is coupled to the uni-junctiontransistor clock at its terminal I50 in series with variable resistanceH51 and diode I152. The frequency of the uni-junction transistor withinclock I40 is arranged to increase in accordance with increased magnitudeof the +e output tachometer voltage impressed upon terminal 150. Thebuss voltage +V also connects to the uni-junction transistor clockcircuit, through potentiometer I56 and series diode 157 as shown. Apresettable shunt capacitor 158 is also in this circuit. The biasingvoltage +V is derived from a three-phase rectifier to which is appliedthreephase voltage E preferably reduced by transformer coupling to a lowlevel for the exemplary five volts for +V The output of rectifier I60comprises a series resistor 161i, shunt filter capacitor I62, and diodeI63 across it. The operation of the traction mode, producing the outputcurves A, B, C per FIG. 17 is now described. Assume first that the motoris not rotating, and the power at buss +V is applied. Such applicationof the power as through contactors etc to power line E, is understood,and also connects the E voltage for rectifier 6G. The clock bias +Vgives the motor a minimum slip frequency, and torque is thereby producedat a low speed, as n-l in FIG. 18. If nothing else were to change, themotor would run along the curve indicated at a, tangent to dashed lineenvelope curve D. However, the motor shaft 146 turns the tachometer I45,raising the DC voltage applied to the clock M0. This directly raises themotor frequency, as will not be understood. The raised clock frequencyprovides the higher speed n-Z corresponding to the curve tangent at b.

This process continues, with higher output tachometer voltages beingapplied to clock 140, which in turn raises the clock frequency and inturn the motor speed, producing the successive curves c, d, e and ftangent to the dashed line D, the latter being at n-6. This increasingspeed and decreasing torque process continues until the load torquerequired in a particular installation keeps the motor from acceleratingfurther. The dashed'line curve D is the resultant torque-speed curvewhich is directly equivalent to that in DC traction motors. Thecomparable curves A, B and C in FIG. I7 are the result of suchtachometer feed-back e to the clock 1M) for respective applied DCvoltages +V indicated. The potentiometer 156 is used for setting theminimum frequency of clock 140, for a selected applied voltage +V. It isthe basic combination effect of voltage +V applied to the motor system,and the generated tachometer M5 voltage e proportional to speed n thatdeter-- change of do buss voltage +V magnitude produces successiveoutput curves as A, B and C. This is similar to the action of appliedvoltage change to a dc series traction motor. It is further noted thatthe dashed-curve loop AA at the left of the A curve peak is typical ofthe system hereof, all nevertheless at high starting torque at the lowspeeds.

The traction operational mode hereof provides curve shapes per A, B, Cof FIG. 17 that are similar to constant horsepower output curves as A,B, C of FIG. 11. However, the traction curves extend operatively all theway down to stall, while significant minimum base speeds are requiredfor operation in the constant horsepower mode. Further, operating speedsin the traction mode depend only upon the mechanical load on the motor,and not on the basic frequency set by the clock (140). High stall torqueis developed at reduced frequency to the motor in the traction mode. Themotor flux densities at stall are relatively high. The product of motorcurrent times flux (l -B) is thus large, resulting in high torque. Thepotentiometer 156 (FIG. 19) is used mainly to set up the system for theminimum frequency appliedto the motor at start up. For example, 1.5hertz would be a practical setting thereof. Also, potentiometer 151 isset for desired regenerative or dynamic braking by the motor system. Itadjusts clock bias from the tachometer in setting for motor accelerationor deceleration corresponding to quadrant (2) operation in FIG. 10.Further, the motor system hereof, in traction mode, can be readilyreversed in direction of rotation through simple logic connection inunit 138 by reversing the d), phase sequence. Such reversal or rotationis not feasible for DC traction series motors.

An important advantage of the traction motor system hereof is that itsdeliverable peak torques are at least ten times the rated torque of themotor. In standard polyphase motors, regardless of frequency or voltageapplied, not much above three times rated torque can be realized. Suchinherent limit of AC polyphase motors is due to their magneticsaturation. After their maximum flux density is reached, no higherdensity can occur, and therefor no higher induced rotor voltage or rotorcurrent accomplished. It is the product of rotor current and gapfluxdensity that produces torque. On the other hand, the quasi-sine steppedwave motor/control system hereof allows fast ddaldt rotor linking ofmagnetic fluxes, even though flux density B be high. With a fairly longrotor time constant, such fast dda/dt linkages allow rotor current to beinduced and continue to flow while the flux density B reaches relativelyhigh levels. Hence, the motor hereof simultaneously experi ences highflux density and high operating current, with peak torque available at alevel that thepolyphase motors cannot match.

What is claimed is:

l. A polyphase motor system comprising a stator and an induction rotor,said stator containing a plurality of half-phase windings in pairedconsequent pole array with one pair for each effective phase for themotor when in polyphase operation, each of said windings being wound infull pitch across 180 magnetic degrees, the complementary windings ofeach said pair being alternated in position along the stator at 180magnetic separation, a controlled rectifier with first and second mainelectrodes for each of said windings, said first main electrodes beinginterconnected, said second main electrodes being connected to theirrespective windings, and control means connected with the controlelectrodes of said controlled rectifiers to selectively establishalternate conduction periods to their associated half-phase windings andprovide a rotating magnetic field in the stator for correspondingoperation of the said induction rotor.

2. A polyphase motor system as claimed in claim 1, in which adjacentturns of the said paired consequent windings are located in common slotsin the stator.

3. A polyphase motor system as claimed in claim 1, in which adjacentturns of the said paired consequent windings are located in common slotsin the stator,,said half-phase windings being wound in the stator insubstantially concentric form;

4. A polyphase motor system as claimed in claim 1, in which each of saidhalf-phase windings provides a plurality of magnetic pole sets, all polesets of which being interrelated in the stator as pairs and in polyphasearray.

5. A polyphase motor system as claimed in claim I, in which saidhalf-phase windings are arrayed about the stator with equal physicalseparation between adjacent windings of each phase pair. A

6. A polyphase motor system as claimed in claim 1, in which saidhalf-phase windings are arrayed about the stator with equal physicalseparation between adjacent windings of each phase pair, associatedwindings of each pair being respectively interconnected into twosymmetrical polyphase winding groups that are 180 apart magnetically inthe stator.

7. A polyphase motor system as claimed in claim I, in which said controlmeans establishes the said conduction periods at respective 180electrical separation between related half-phase pairs to provide saidrotating magnetic field in the stator.

8. A polyphase motor system as claimed in claim 6, in which said controlmeans establishes the said controlled rectifier conduction periods atrespective l electrical separation between related half-phase pairs insaid winding groups to provide said rotating magnetic field in thestator.

9. A polyphase motor system as claimed in claim 1,

in which said half-phase windings are arrayed in the stator withrelatively low mutual magnetic induction between the windings ofrespective pairs, whereby magnetic interaction among said statorwindings cyclically effects self-clearing of the controlled rectifierconductions should any extend beyond its predetermined conductionperiod. 10. A polyphase motor system as claimed in claim 6, in whichsaid control means includes a timing circuit that effects saidsuccessive conduction periods in equally spaced time intervals amongsaid half-phase windings once during each cycle of applied motorfrequency, said timing circuit having an electronic clock, and circuitmeans coupled to said clock and arranged to control its frequency andthereby the duration of said conduction .periods to effect apredetermined mode of operation of the induction rotor.

11. A polyphase motor system as claimed in claim 7, in which saidcontrol means includes a timing circuit that effects said successiveconduction periods in equally spaced time intervals among saidhalf-phase windings once during each cycle of applied motor frequency,said timing circuit having an electronic clock, and circuit meanscoupled to said clock and arranged

1. A polyphase motor system comprising a stator and an induction rotor, said stator containing a plurality of half-phase windings in paired consequent pole array with one pair for each effective phase for the motor when in polyphase operation, each of said windings being wound in full pitch across 180 magnetic degrees, the complementary windings of each said pair being alternated in position along the stator at 180* magnetic separation, a controlled rectifier with first and second main electrodes for each of said windings, said first main electrodes being interconnected, said second main electrodes being connected to their respective windings, and control means connected with the control electrodes of said controlled rectifiers to selectively establish alternate conduction periods to their associated halfphase windings and provide a rotating magnetic field in the stator for corresponding operation of the said induction rotor.
 2. A polyphase motor system as claimed in claim 1, in which adjacent turns of the said paired consequent windings are locAted in common slots in the stator.
 3. A polyphase motor system as claimed in claim 1, in which adjacent turns of the said paired consequent windings are located in common slots in the stator, said half-phase windings being wound in the stator in substantially concentric form.
 4. A polyphase motor system as claimed in claim 1, in which each of said half-phase windings provides a plurality of magnetic pole sets, all pole sets of which being interrelated in the stator as pairs and in polyphase array.
 5. A polyphase motor system as claimed in claim 1, in which said half-phase windings are arrayed about the stator with equal physical separation between adjacent windings of each phase pair.
 6. A polyphase motor system as claimed in claim 1, in which said half-phase windings are arrayed about the stator with equal physical separation between adjacent windings of each phase pair, associated windings of each pair being respectively interconnected into two symmetrical polyphase winding groups that are 180* apart magnetically in the stator.
 7. A polyphase motor system as claimed in claim 1, in which said control means establishes the said conduction periods at respective 180* electrical separation between related half-phase pairs to provide said rotating magnetic field in the stator.
 8. A polyphase motor system as claimed in claim 6, in which said control means establishes the said controlled rectifier conduction periods at respective 180* electrical separation between related half-phase pairs in said winding groups to provide said rotating magnetic field in the stator.
 9. A polyphase motor system as claimed in claim 1, in which said half-phase windings are arrayed in the stator with relatively low mutual magnetic induction between the windings of respective pairs, whereby magnetic interaction among said stator windings cyclically effects self-clearing of the controlled rectifier conductions should any extend beyond its predetermined conduction period.
 10. A polyphase motor system as claimed in claim 6, in which said control means includes a timing circuit that effects said successive conduction periods in equally spaced time intervals among said half-phase windings once during each cycle of applied motor frequency, said timing circuit having an electronic clock, and circuit means coupled to said clock and arranged to control its frequency and thereby the duration of said conduction periods to effect a predetermined mode of operation of the induction rotor.
 11. A polyphase motor system as claimed in claim 7, in which said control means includes a timing circuit that effects said successive conduction periods in equally spaced time intervals among said half-phase windings once during each cycle of applied motor frequency, said timing circuit having an electronic clock, and circuit means coupled to said clock and arranged to control its frequency and thereby the duration of said conduction periods to effect a predetermined mode of operation of the induction rotor.
 12. A polyphase motor system as claimed in claim 11, further including terminal means for connecting said half-phase windings to a source of DC operating voltage, said circuit means being arranged to vary the frequency of said clock and effect substantially constant horsepower output operation of the rotor.
 13. A polyphase motor system as claimed in claim 11, further including electrical means providing a variable source of operating DC voltage for said half-phase windings, said circuit means including a Zener diode, said variable DC voltage being coupled to said Zener diode and thereby to said clock, whereby variation of the applied dc voltage directly effects substantially constant torque output by the rotor. 